Ultrasonic flow meter with subsampling of ultrasonic transducer signals

ABSTRACT

An ultrasonic flow meter is disclosed, including a switching unit for switching electrical transmission signals between a signal generator and at least two ultrasonic transducers and for switching electrical reception signal between the transducers and a receiver circuit, wherein the switching unit is coupled to an output terminal of an operational amplifier of the signal generator and to an inverting input terminal of an operational amplifier of the receiver circuit. Furthermore, a method for characterizing an ultrasonic transducer is disclosed, including the step of determining directly from one or more supply current signals for an active component of a signal generator one or more quantities useful for characterizing the transducer. Furthermore, a method for determining the time delay of an ultrasonic signal in a flow path of an ultrasonic flow meter is disclosed, including the step of comparing physically transmitted, delayed and received signals with simulated non-delayed signals.

FIELD OF THE INVENTION

The present invention relates to an ultrasonic flow meter for measuringthe flow of a fluid, in particular within the field of transit time flowmetering.

BACKGROUND OF THE INVENTION

Generally speaking, flow metering by means of the transit time methodincludes placing two ultrasonic transducers with a suitable mutualdistance in the flow path in which the flow of a fluid is to bemeasured. An ultrasound signal, typically of a frequency of a fewmegahertz and a duration of a few microseconds, is transmitted throughthe fluid from the first transducer to the second transducer, and afirst transmit time is recorded. Next, a similar ultrasound signal istransmitted through the fluid in the opposite direction, i.e. from thesecond transducer to the first transducer, and a second transmit time isrecorded. Knowing the physical distance between the two transducers, thedifference between the two recorded transmit times can be used forcalculating the flow rate of the fluid flowing in the flow path.However, the calculated flow rate must be corrected by means of acorrection table taking into account the sound velocity and viscosity ofthe fluid. Both of those characteristics being dependent on thetemperature, a correction table with correction values depending on thetemperature is sufficient when the type of fluid is known.

One problem to be faced when working with this type of flow meters isthat the transducer parameters are not only very likely to differbetween samples but also change over time and when the temperaturechanges. Such differences and changes alter the shape of the receivedsignal, making it difficult to use this shape as basis for thecalculation of the absolute transit time.

During the last 25 years, ultrasonic flow metering has seen a dramaticdevelopment from low volume laboratory instruments to standard equipmentproduced in very high volume. Technical and commercial challenges havebeen overcome to a degree that the technology is now competitive againstmost other methods including mechanical meters in many areas of flowmetering. For instance, highly accurate flow meters produced in highvolumes are now commonly used as water meters, heat meters, gas metersand other meters used for billing.

Some of the challenges still to be worked on are improving the meters sothat they are less sensitive to electrical and acoustical noise whilestill keeping the meters stable and producible and without sacrificingcost and power consumption. Sensitivity to noise can be decreased byincreasing the signal to noise ratio, the most effective method beingincreasing the signal.

Typical acoustical noise sources in an ultrasonic flow meter are edgesin the flow current and external vibrations, both producing a fixedacoustical noise level independent of the ultrasound generated by themeter itself. The sensitivity to the acoustical noise can be reduced byincreasing the acoustical signal produced by the transducers or bychanging the physical shape of the flow meter.

Electrical noise in an ultrasonic flow meter has many sources, such asthermal noise, externally induced (by electromagnetic, electric ormagnetic fields or by wire) voltages and currents, or internally induced(from other signals or clocks in the electric circuit) cross coupling,some of which are signal level dependent and some of which areindependent of the signal level. The most effective way to reduce thesensitivity to electrical noise is by increasing the electrical signalsinvolved and by keeping impedances of electrical nodes as low aspossible in order to reduce the influence of the sources of electricalnoise.

Many different electrical circuits relating to these subjects are knownin the art, such as GB 2 017 914 (Hemp), U.S. Pat. No. 4,227,407(Drost), DE 196 13 311 (Gaugler), U.S. Pat. No. 6,829,948 (Nakabayashi),EP 0 846 936 (Tonnes) and EP 1 438 551 (Jespersen), each havingstrengths and weaknesses.

The two last-mentioned documents (Tonnes and Jespersen) show transducercouplings having the benefit that the impedance as seen from thetransducers is the same in the transmit situation and in the receivesituation. Discussions in the two patent documents explain that thisfeature is a prerequisite for the whole flow meter to demonstratestability and producibility in real life situations, i.e. withoutunrealistic requirements on matching between components in the meter.The reason for this fact is that the exact impedance match allows theflow meter to fully exploit the reciprocity theorem.

Although the connection between reciprocity and stable flow metering hasbeen known for many years, the couplings shown in these patent documentsare the only practical ways, known to date, that fully achieve absorbingthe natural tolerances of piezoelectric ultrasonic transducers so thatproducible and stable flow meters can be produced.

The transducer couplings shown in both of these two documents comprisean impedance, which has the function of converting the current signalreceived from the transducer to a measurable voltage signal.Unfortunately, as explained in further detail below, this impedance alsolimits the electrical signal that can be supplied to the transducers,and in order to produce in the largest possible received voltage signal,the size of the impedance is restricted to be in the range between 0.5and 2 times the impedance of the ultrasonic transducers at the frequencyof interest.

Nakabayashi (U.S. Pat. No. 6,829,948) has another approach, in which thegenerator and the receiver are implemented by two different means, butin this configuration, the received signal strength is sacrificed forstable results at changing transducer parameters.

It is an object of the present invention, which is described in thefollowing, to overcome the above-identified problems and to provide astable, producible flow meter, which is capable of transmitting a highacoustical signal.

SUMMARY OF THE INVENTION

The present invention relates to a method for determining the absolutetransit time of an ultrasonic signal in a flow path of an ultrasonicflow meter comprising two ultrasonic transducers, said method comprisingthe steps of:

-   -   monitoring a current to an active component of a signal        generator driving a transducer from one or more voltage supplies        during and after a feeding of an input signal to the signal        generator, thus obtaining a supply current signal for the        transducer,    -   emulating a flow meter response similar to an output signal from        a receiver circuit of the flow meter as the output signal would        be, if there was no time delay in the transmission of an        ultrasonic signal between the two ultrasonic transducers,    -   comparing the emulated flow meter response to a measured flow        meter response actually received by the receiver circuit, and    -   calculating the absolute transit time as the time difference        between the emulated flow meter response and the measured flow        meter response.

Such a method has shown to be efficient and result in very precisedetermination of the absolute transit time as compared to previouslyknown methods.

In an embodiment of the invention, the step of emulating a flow meterresponse comprises:

-   -   feeding a single pulse input signal to a signal generator        driving a transducer, said signal generator comprising an active        component,    -   monitoring the current to the active component from one or more        voltage supplies during and after the feeding of the input        signal to the signal generator, thus obtaining a single pulse        supply current signal for the transducer,    -   adjusting the single pulse supply current signal and for        obtaining an emulated single pulse response of the transducer,    -   repeating the three previous steps for the other transducer,        thus obtaining another emulated single pulse response,    -   finding the single pulse response of the system by making a        convolution of the two obtained single pulse responses of the        transducers, and    -   calculating the emulated flow meter response by combining a        number of instances of the found single pulse response of the        system, which are repeated with suitable delays.

This has shown to be an efficient way of obtaining emulated flow meterresponses very similar to the measured flow meter responses.

In an embodiment of the invention, the step of emulating a flow meterresponse comprises:

-   -   feeding a pulsating input signal to a signal generator driving a        transducer, said signal generator comprising an active        component,    -   monitoring the current to the active component from one or more        voltage supplies during and after the feeding of the input        signal to the signal generator, thus obtaining one or more        supply current signal,    -   determining directly from the obtained one or more supply        current signal or from one or more resulting signals derived        from the one or more obtained supply current signal one or more        quantities useful for characterizing the transducer,    -   repeating the three previous steps for the other transducer,        thus obtaining similar quantities useful for characterizing the        other transducer,    -   using the determined characteristic quantities of the        transducers for finding an equivalence model of the transducers        and establishing a numerical simulation model of the transducers        and electronic circuits of a signal generator and/or a receiver        circuit of the flow meter, and    -   simulating the flow meter system by entering an input signal        function or a sampled version of a physical transmission signal        reaching the first transducer into the numerical simulation        model, thereby obtaining the emulated flow meter response.

This is another efficient way of obtaining emulated flow meter responsesvery similar to the measured flow meter responses.

In an embodiment of the invention, the quantities useful forcharacterizing the transducer include an oscillation period and/or adamping coefficient determined from at least a part of one or more ofthe obtained signals and/or the derived signals, said signal partrepresenting a dampened oscillation.

The oscillation period and the damping coefficient relating to adampened oscillation of the transducer are very useful characteristicsof the transducers, which are very suitable for constructing an adequateequivalence model of the transducers.

In an embodiment of the invention, the one or more supply currentsignals are obtained by monitoring the voltage across one or morecurrent sensing resistors being arranged in series between the activecomponent of the signal generator and one or more voltage supplies ofthe signal generator.

This is a simple, stable and well-known method for measuring a currentsignal.

In an embodiment of the invention, the step of calculating the absolutetransit time comprises:

-   -   transforming the emulated flow meter response and the measured        flow meter response into the frequency domain, for instance by        means of Fast Fourier Transformation,    -   determining the phase angle between the two flow meter responses        in the frequency domain for at least two different frequencies,        and    -   determining the absolute transit time by calculating the group        time delay from the two determined phase angles.

The use of Fast Fourier Transformation and working in the frequencydomain substantially reduces the amount of calculation needed todetermine the absolute transit time.

In an embodiment of the invention, the step of calculating the absolutetransit time comprises:

-   -   finding filtered envelopes of the emulated flow meter response        and the measured flow meter response, respectively,    -   identifying the two points in time, in which the filtered        envelopes have reached 50% of their maximum value, respectively,        and    -   calculating the absolute transit time as the time difference        between the two identified points in time.

This method has shown to provide a very precise determination of theabsolute transit time of the ultrasonic signal passing through the flowpath of the flow meter.

In an aspect of the invention, it relates to an ultrasonic flow metercomprising at least one ultrasonic transducer and a signal generator forgenerating electrical signals to the transducer, the signal generatorcomprising an active component, wherein the flow meter further includesmeans for measuring one or more power supply currents to the activecomponent of the signal generator.

This enables for the possibility of characterising the transducers whilethey are arranged in the flow meter.

In an embodiment of the invention, the means for measuring one or morepower supply currents comprise a resistor inserted in series between thesource of the positive supply voltage and the active component.

This is a simple, stable and well-known method for measuring a currentsignal.

In an embodiment of the invention, the means for measuring one or morepower supply currents comprise a resistor inserted in series between thesource of the negative supply voltage and the active component.

Measuring the supply currents in both power supply connections enablesfor a faster characterization of the transducers, measuring two supplycurrent signals simultaneously.

In an aspect of the invention, it relates to a method for characterizingan ultrasonic transducer, said method comprising the steps of:

-   -   feeding a pulsating input signal to a signal generator driving        the transducer, said signal generator comprising an active        component,    -   monitoring the current to the active component from one or more        voltage supplies during and after the feeding of the input        signal to the signal generator, thus obtaining one or more        supply current signal, and    -   determining directly from the obtained one or more supply        current signal or from one or more resulting signals derived        from the one or more obtained supply current signal one or more        quantities useful for characterizing the transducer.

This method enables for characterization of the transducers while theyare arranged in the flow meter.

In an embodiment of the invention, the active component is anoperational amplifier.

In another embodiment of the invention, the active component is adigital circuit driving the transducer.

This reflects that different types of active components can be used inthe signal generator.

In an embodiment of the invention, the one or more supply currentsignals are obtained by monitoring the voltage across one or morecurrent sensing resistors being arranged in series between the activecomponent of the signal generator and one or more voltage supplies ofthe signal generator.

This is a simple, stable and well-known method for measuring a currentsignal.

In an embodiment of the invention, the quantities useful forcharacterizing the transducer include an oscillation period and/or adamping coefficient determined from at least a part of one or more ofthe obtained signals and/or the derived signals, said signal partrepresenting a dampened oscillation.

The oscillation period and the damping coefficient relating to adampened oscillation of the transducer are very useful characteristicsof the transducers, which are very suitable for constructing an adequateequivalence model of the transducers.

In an embodiment of the invention, it relates to a method fordetermining the time delay of the ultrasonic signal in the flow path ofan ultrasonic flow meter, said method comprising the steps of:

-   -   characterizing two transducers of the flow meter by determining        characteristic quantities, such as the angular frequency and the        damping coefficient of dampened oscillations of the transducers,    -   using the determined characteristic quantities of the        transducers for finding an equivalence model of the transducers        and establishing a numerical simulation model of the transducers        and electronic circuits of a signal generator and/or a receiver        circuit of the flow meter,    -   simulating the flow meter system by entering an input signal        function or a sampled version of a physical transmission signal        reaching the first transducer into the numerical simulation        model, thereby obtaining a simulation model response        corresponding to the output signal from the receiver circuit as        it would be according to the model, if there was no time delay        in the transmission of the ultrasonic signal between the two        transducers,    -   recording the physical flow meter response actually received by        the receiver circuit, and    -   calculating the absolute transit time by determining the time        delay of the physical flow meter response as compared to the        simulation model response.

This method has shown to provide a very precise determination of theabsolute transit time of the ultrasonic signal passing through the flowpath of the flow meter.

In an embodiment of the invention, the step of calculating the absolutetransit time comprises the steps of:

-   -   finding filtered envelopes of the simulation model response and        the physical flow meter response, respectively,    -   identifying the two points in time, in which the filtered        envelopes have reached 50% of their maximum value, respectively,        and    -   calculating the absolute transit time as the time difference        between the two identified points in time.

This method of calculating the absolute transit time has shown to bevery precise and reproducible, taking into account that the transducerparameters not only are very likely to differ between samples but alsochange over time and when the temperature changes.

In an aspect of the invention, it relates to an ultrasonic flow metercomprising at least one ultrasonic transducer and a signal processingunit for processing electrical signals received from the at least oneultrasonic transducer, wherein the signal processing unit is arranged todigitalize a continuous signal at a sample frequency of less than twicethe resonance frequency of the at least one ultrasonic transducer.

This allows for the use of slower and cheaper analogue-digitalconverters in the flow meter than would otherwise be needed.

In an aspect of the invention, it relates to an ultrasonic flow metercomprising a flow path for fluid flow, at least two ultrasonictransducers acoustically coupled to the flow path, the one transducerarranged upstream of the other transducer along the flow path, a signalgenerator for generating electrical transmission signals to thetransducers, the signal generator comprising a negative feedback coupledoperational amplifier, a receiver circuit for receiving electricalreception signals from the transducers, the receiver circuit comprisinga negative feedback coupled operational amplifier, a switching unit forswitching electrical transmission signals between the signal generatorand the transducers and for switching electrical reception signalbetween the transducers and the receiver circuit, a signal processingunit for providing an output indicative of the flow rate in the flowpath based on the electrical reception signals, wherein the switchingunit is coupled to an output terminal of the operational amplifier ofthe signal generator and the switching unit is coupled to an invertinginput terminal of the operational amplifier of the receiver circuit.

Thus, the invention relates especially to high accuracy, high volume,low power and low cost consumption meters for billing purposes.

By coupling the switching unit, and thereby the transducers, to theoutput terminal of an operational amplifier of the signal generator andto the inverting input terminal of the operational amplifier of thereceiver circuit, it is achieved that the impedance as seen from thetransducers is the same regardless of whether the transducers areoperated as transmitters or as receivers. This means that thereciprocity theorem for linear passive circuits applies for the flowmeter, which is important for its stability and producibility.

In an embodiment of the invention, an output impedance of the signalgenerator and an input impedance of the receiver circuit are negligiblecompared to the impedance of the transducers, such as less than 10 Ohms,preferably less than 1 Ohm, most preferred less than 0.1 Ohm.

Choosing a very low output impedance of the signal generator SG and avery low input resistance of the receiver circuit RC is advantageous forobtaining that the transducers will experience the two impedances asbeing sufficiently close to each other, and for assuring that theattenuation of the electrical transmission and reception signals isminimized.

In an embodiment of the invention, the signal generator and the receivercircuit share at least one active component.

This is advantageous for saving component costs when producing the flowmeter.

In an embodiment of the invention, all active components of the signalgenerator are completely separate from all active components of thereceiver circuit.

Using different active components for the signal generator and for thereceiver circuit enables for the possibility that the signal can betransmitted all the way through the flow meter without any switchingover of the transmission path having to take place during thetransmission.

In an embodiment of the invention, one or more of the operationalamplifiers in the signal generator and the receiver circuit are currentfeedback operational amplifiers.

The use of current feedback amplifiers is beneficial, because suchamplifiers have a lower input impedance on the inverting input terminaland a higher bandwidth and a lower power consumption and a higher gainat high frequencies than other types of operational amplifiers.

In an embodiment of the invention, one or more of the operationalamplifiers in the signal generator and the receiver circuit are operatedat an input common mode voltage, the AC component of which issubstantially zero or at least negligible.

This is of importance for some types of operational amplifiers,especially the fastest ones with the highest bandwidth and very lowcurrent consumption, as the voltage swing allowed on the input forlinear operation can be very limited.

In an embodiment of the invention, the at least two transducers arearranged to be able to transmit an ultrasound signal simultaneously.

In this configuration, the electrical transmission signal is only to besent half as many times as in other configurations, saving battery lifetime. In addition, the transit time is measured simultaneously in thetwo directions and, thus, no sudden change of the flow velocity betweenmeasurements in the two directions can occur.

BRIEF DESCRIPTION OF THE DRAWINGS

In the following, a few exemplary embodiments of the invention aredescribed and explained in more detail with reference to the drawings,where

FIG. 1 illustrates schematically the overall structure of an ultrasonicflow meter for transit time flow metering as known in the art,

FIG. 2a illustrates schematically a coupling of ultrasonic transducersin an ultrasonic flow meter known in the art,

FIG. 2b illustrates schematically a coupling of ultrasonic transducersin another ultrasonic flow meter known in the art,

FIG. 3 illustrates schematically a coupling of ultrasonic transducers inan ultrasonic flow meter according to an embodiment of the invention,

FIG. 4 illustrates schematically a coupling of ultrasonic transducers inan ultrasonic flow meter according to another embodiment of theinvention,

FIG. 5 illustrates schematically a coupling of ultrasonic transducers inan ultrasonic flow meter according to yet another embodiment of theinvention,

FIG. 6 shows a diagram of most of the essential electronic components inan ultrasonic flow meter according to an embodiment of the invention,

FIG. 7a illustrates schematically a set-up for performing a first stepin obtaining a supply current signal of an active component driving anultrasonic transducer,

FIG. 7b illustrates schematically a set-up for performing a second stepin obtaining a supply current signal of an active component driving anultrasonic transducer,

FIG. 7c illustrates an advantageous way of connecting a signal generatorand a receiver circuit for obtaining such supply current signals,

FIG. 8a illustrates schematically a first step in obtainingcharacteristics of an ultrasonic transducer from supply current signalsof an active component driving the transducer,

FIG. 8b illustrates schematically a second step in obtainingcharacteristics of an ultrasonic transducer from supply current signalsof an active component driving the transducer,

FIG. 9 illustrates a well-known equivalence diagram of an ultrasonictransducer,

FIG. 10 illustrates some of the steps in the derivation of a simpleequivalence diagram of an ultrasound flow meter according to theinvention,

FIG. 11 illustrates the differences and similarities between a physicaland a simulated signal chain through a flow meter according to theinvention,

FIG. 12 illustrates schematically a method according to the inventionfor determining very precisely the time delay of the ultrasonic signalin the flow path,

FIG. 13 illustrates schematically in more detail a method forcalculation of the absolute transit time from response signals obtainedby a method as the one illustrated in FIG. 12,

FIG. 14 illustrates an equivalence diagram of a signal generator and anultrasonic transducer connected thereto according to the invention,

FIG. 15a illustrates a single pulse signal,

FIG. 15b illustrates a supply current signal for a first ultrasoundtransducer obtained in response to the single pulse in FIG. 15 a,

FIG. 15c illustrates a supply current signal for a second ultrasoundtransducer obtained in response to the single pulse in FIG. 15 a,

FIG. 15d illustrates a supply current signal obtained without anyultrasound transducers in response to the single pulse in FIG. 15 a,

FIG. 16a illustrates a calculated single pulse response of a firstultrasound transducers,

FIG. 16b illustrates a calculated single pulse response of a secondultrasound transducers,

FIG. 16c illustrates a calculated single pulse response of a completeultrasound transducer system according to the invention,

FIG. 17a illustrates an emulated flow meter response,

FIG. 17b illustrates a measured flow meter response,

FIG. 18a illustrates the theoretical relation between the magnitudes ofthe real and the emulated flow meter responses in the frequency domain,

FIG. 18b illustrates the theoretical phase angles between the real andthe emulated flow meter responses in the frequency domain,

FIG. 19 illustrates an example of corresponding emulated and measuredflow meter responses,

FIG. 20a illustrates the magnitude in the frequency domain of theemulated flow meter response shown in FIG. 19,

FIG. 20b illustrates the magnitude in the frequency domain of themeasured flow meter response shown in FIG. 19,

FIG. 21a illustrates the actual phase angles between the flow meterresponses shown in FIG. 19,

FIG. 21b illustrates a segment of the graph shown in FIG. 21 a,

FIG. 22 illustrates the slopes of the segment shown in FIG. 21 b,

FIG. 23a illustrates an example of the frequency spectrum of acontinuous signal,

FIG. 23b illustrates schematically the spectral consequences ofundersampling the continuous signal, whose frequency spectrum isillustrated in FIG. 23 a,

FIG. 23c illustrates schematically how the undersampled signal can bereconstructed by changing the sampling frequency and filtering thesignal,

FIG. 24a illustrates an example of a continuous signal,

FIG. 24b illustrates the digital samples obtained by sampling the signalof FIG. 24a at a sampling frequency of ⅚ of the signal frequency,

FIG. 25a illustrates the reconstruction of the signal of FIG. 24a usinga wide band FIR reconstruction filter,

FIG. 25b illustrates the reconstruction of the same signal using anarrow band FIR reconstruction filter,

FIG. 26 illustrates schematically a method for finding the amplitude andphase of an undersampled continuous signal,

FIG. 27 illustrates schematically a method for reconstructing anundersampled continuous signal without distortion, and

FIG. 28 illustrates schematically an extended method according to theinvention for determining very precisely the time delay of theultrasonic signal in the flow path.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 shows the overall structure of an ultrasonic flow meter fortransit time flow metering as known in the art. A controller unit CUcontrols the operation of a signal generator SG, a switching unit SU anda receiver circuit RC, where the switching unit SU sets up differentelectrical connections between the signal generator SG and the receivercircuit RC on one side and two ultrasonic transducers TR1, TR2 on theother side. The two transducers TR1, TR2 are arranged in a flow path FP,in which a fluid flow is to be metered, one transducer TR1 upstream ofthe other transducer TR2 along the flow path FP.

In principle, the flow metering is performed in three steps:

-   -   1. The switching unit SU is set up to connect the signal        generator SG to the first transducer TR1 and to connect the        second transducer TR2 to the receiver circuit RC.        -   An electrical transmission signal, typically a pulsating            signal of a frequency of a few megahertz and duration of a            few microseconds, is sent from the signal generator SG            through the switching unit SU to the first transducer TR1,            from which the signal is transmitted as an ultrasonic signal            through the fluid to the second transducer TR2. From TR2,            the signal continues as an electrical current reception            signal through the switching unit SU to the receiver circuit            RC, in which the reception signal is converted to a voltage            signal.        -   A signal processing unit (part of the controller unit CU in            the configuration shown in FIG. 1), analyses the voltage            signal, calculates the transit time of the ultrasound signal            through the fluid from transducer TR1 to transducer TR2 from            the delay of the electrical reception signal as compared to            the electrical transmission signal, and records this transit            time (t₁).    -   2. The configuration of the switching unit SU is changed to        connect the signal generator SG to the second transducer TR2 and        to connect the first transducer TR1 to the receiver circuit RC.        -   An electrical transmission signal is sent from the signal            generator SG and an electrical reception signal is received            by the receiver circuit RC as in step 1, only this time the            ultrasonic signal is transmitted through the fluid in the            opposite direction, i.e. from the second transducer TR2 to            the first transducer TR1.        -   Again, the signal processing unit calculates and records the            transit time (t₂).    -   3. The signal processing unit calculates an indication of the        flow in the flow path FP from a formula of the form:

$\begin{matrix}{\Phi \propto {{K\left( {{t_{1} - t_{2}},{t_{1} + t_{2}}} \right)} \cdot \frac{t_{1} - t_{2}}{\left( {t_{1} + t_{2}} \right)^{2}}}} & \left( {{Equation}\mspace{14mu} 1} \right)\end{matrix}$

-   -   -   where Φ is the flow indication, which is proportional to the            fraction shown in Equation 1 multiplied by a correction            factor K found in a table of correction factors, which is            determined once and for all for a given type of flow meter            for a given fluid.        -   This table of correction factors takes into account a number            of physical quantities, such as, for instance, the            dimensions and physical configuration of the flow path FP in            the flow meter and the viscosity of the fluid.

As can be seen from Equation 1, once the table of correction factors hasbeen established, the flow indication can be calculated from the twoquantities (t₁−t₂) and (t₁+t₂).

The first of these quantities, (t₁−t₂), which is the difference betweenthe two transit times, is typically in the order of a few nanoseconds,but can easily be determined by finding the phase difference between thetwo reception signals. This can be done very precisely (with an accuracyof down to between 10 and 100 picoseconds) by several analogue anddigital methods well-known through many years, due to the fact that thetwo reception signals are identical except for a phase difference due tothe different transit times (t₁ and t₂), given that the reciprocitytheorem for linear passive circuits applies. Generally, this is the caseif it is assured that the impedance, as seen from the transducers TR1,TR2 is the same, regardless of whether the transducers TR1, TR2 areacting as transmitters or receivers of ultrasound.

On the other hand, it is very difficult to calculate accurately theother quantity, (t₁+t₂), which is the sum of the two transit times,typically in the order of a few microseconds, because it involves acalculation of the exact transit times (t₁ and t₂), which again requiresa very precise determination of the front edge of each of the receptionsignals, which is by no means a simple task due to the shape of thereception signals.

Therefore, in many known flow meters, this quantity is, in fact, notcalculated. Instead, it is estimated using the following equation:

$\begin{matrix}{\frac{d}{c} = {t \approx t_{1} \approx t_{2}}} & \left( {{Equation}\mspace{14mu} 2} \right)\end{matrix}$

In this equation, d is the distance between the two transducers TR1, TR2and c is the velocity of ultrasound in the actual fluid, the flow ofwhich is being metered. For a given flow meter, d is known from thephysical positions of the transducers TR1, TR2 in the flow path FP, and,for a given temperature, the velocity of ultrasound in a given fluid canbe found in a table. Thus, by measuring the temperature of the fluid, anestimate of t₁ and t₂ can be found, which can then be used forestimating the quantity (t₁+t₂) to be used in Equation 1.

FIGS. 2a and 2b illustrate schematically examples of couplings ofultrasonic transducers in ultrasonic flow meters according to theinventions of Tonnes (EP 0 846 936) and Jespersen (EP 1 438 551),respectively.

Implemented correctly, both couplings assure that the reciprocitytheorem for linear passive circuits applies.

In the coupling shown in FIG. 2a , the electrical transmission signal istransmitted from the signal generator SG to transducer TR1 through thesignal impedance Zsig. In the coupling shown in FIG. 2b , on the otherhand, the signal generator SG comprises a negative feedback coupledamplifier circuit, wherein a digital pulsating signal is connected tothe non-inverting input terminal of an operational amplifier OPsg andthe signal impedance Zsig forms a feedback between the output terminaland the inverting input of the operational amplifier OPsg. TransducerTR1 is connected to the inverting input terminal of the operationalamplifier OPsg through an adaptation impedance Zad, which is muchsmaller than the signal impedance Zsig and therefore, in practice,negligible.

In both of the shown couplings, the switching unit SU comprises twoswitches SW1, SW2 arranged to be able to connect the two transducersTR1, TR2, respectively to a common conductor CC, which connects thesignal generator SG to the receiver circuit RC. In both cases, theposition of each of the switches SW1, SW2 has to be changed during theflow metering in order to assure that, at the time of transmission ofthe electrical transmission signal from the signal generator SG, one ofthe transducers TR1, TR2 is connected to the common conductor CC,whereas, at the time of reception of the electrical reception signal bythe receiver circuit RC, the other transducer TR2, TR1 is connected tothe common conductor CC. This change of switch positions must take placeafter the ultrasonic signal has left the transmitting transducer TR1,TR2, but before it reaches the receiving transducer TR2, TR1. Thus, thetiming is very crucial.

The signal impedance Zsig through which the signal current will run inboth of the couplings shown in FIGS. 2a and 2b has the function ofconverting the current signals received from the transducers TR2, TR1 tomeasurable voltage signals.

The size of the voltage signals is found by multiplying the receivedcurrents by the signal impedance Zsig, a large signal impedance Zsigresulting in a large received voltage signal.

Unfortunately, due to practical limitations on the supply voltage to thesignal generator SG, the signal impedance Zsig also limits theelectrical signal that can be supplied to the transducers TR1, TR2,because the signal impedance Zsig is also present during thetransmission of a signal to the transducers TR1, TR2. Thus, the outputvoltage from the signal generator SG has to be larger than the signalrequested on the transducers TR1, TR2. The compromise resulting in thelargest received voltage signal is a value of the signal impedance Zsigin the range between 0.5 and 2 times the impedance of the ultrasonictransducers TR1, TR2 at the frequency of interest.

The present invention, on the other hand, provides a stable, producibleflow meter, which is capable of transmitting a high acoustical signal,at the same time amplifying the received current signal by a high signalimpedance and having a low impedance at sensitive nodes in thecircuitry.

The basic idea in the present invention is to connect the transducersTR1, TR2 to different nodes in the transmit and receive situation makingsure that the reciprocity theorem for linear passive circuits stillapplies, i.e. without sacrificing the characteristic that the impedanceas seen from the transducer TR1, TR2 is the same regardless of whetherthe transducer TR1, TR2 is operated as a transmitter or as a receiver.

This is achieved by coupling the switches SW1, SW2 of the switching unitSU to the output terminal of an operational amplifier OPsg of the signalgenerator SG and to the inverting input terminal of the operationalamplifier OPrc of the receiver circuit RC as illustrated schematicallyin FIG. 3, which shows an embodiment of the present invention.

By using an operational amplifier OPsg with a very low output impedancein the signal generator SG and by choosing feedback components resultingin appropriate feedback impedances Zfb,sg, Zfb,rc for constructingnegative feedback circuits for the two operational amplifiers OPsg,OPrc, respectively, it is possible to obtain a signal generator SG witha very low output impedance and a receiver circuit RC with a very lowinput impedance, at the same time accounting for the parasiticcomponents of the operational amplifiers OPsg, OPrc. The very lowimpedances are obtained by coupling operational amplifiers with a veryhigh gain at the frequency of interest with a negative feedback.

Choosing a very low output impedance of the signal generator SG and avery low input resistance of the receiver circuit RC is advantageous forat least four reasons:

First of all, if these impedances are both sufficiently low as comparedto the impedances of the transducers TR1, TR2, and the parasiticcomponents are accounted for, then, even though there may actually be aminimal difference between the output impedance of the signal generatorSG and the input impedance of the receiver circuit RC, the transducersTR1, TR2 will experience the two impedances as being sufficiently closeto each other. This means that, substantially, the reciprocity theoremfor linear passive circuits applies, and the flow meter is stable andproducible.

For a proper implementation, the output impedance of the signalgenerator SG and the input impedance of the receiver circuit RC shouldboth be negligible compared to these transducer impedances, i.e. lessthan 1%, preferably less than 0.1%, of the transducer impedances.Depending on the transducer size and material and on the frequency ofthe transmitted signal, the transducer impedances at the frequencies ofinterest normally fall within the range from 100 Ohms to 1000 Ohms.

Secondly, the choice of small output and input impedances assures thatthe attenuation of the electrical transmission and reception signals isminimized, maximizing the output signal received by the receiver circuitRC.

Thirdly, very low impedances are chosen because mid-range values can behard to match within negligible tolerances in different parts of thecircuits, especially as it is the complex impedance and not just theabsolute resistance value that has to be taken into account.

Fourthly, a low circuit impedance is less susceptible to interferencefrom external noise sources.

An obvious advantage of the coupling shown in FIG. 3 is that by usingtwo different operational amplifiers OPsg, OPrc in the signal generatorSG and in the receiver circuit RC, respectively, no switching isnecessary between the transmission and the reception of the ultrasonicsignal between the two transducers TR1, TR2. When the switches SW1, SW2are in the positions as shown in FIG. 3, the electrical transmissionsignal will pass from the signal generator SG through the second switchSW2 to the second transducer TR2, from which the ultrasonic signal willbe transmitted to the first transducer TR1 through the flow path (FP),whereupon the electrical reception signal will reach the receivercircuit RC from the first transducer TR1 through the first switch SW1.In order to transmit the ultrasonic signal in the opposite direction,the positions of both switches SW1, SW2 must be inverted, and again thesignal can be transmitted all the way from the signal generator SG tothe receiver circuit RC without any switching taking place during thetransmission. In this way, switching noise is avoided and a moreaccurate metering may be performed.

It is known from the art that separate circuits have been used forconstructing the signal generator and the receiver circuit. In thesecases, however, either the transducers experience different impedancesin transmit and receive situations, the amplification factors aredifferent for the two transducers, the invention is not sufficientlydisclosed to be more than theoretical, or a high impedance has beenchosen.

The latter has the disadvantage that designing a signal generator havingan output impedance very much (100 to 1000 times) larger than thetransducer impedances at the frequencies of interest (100 kHz to 10 MHz)is very challenging. It also has the disadvantage that the optimizationof the output signal amplitude has to take into account the transducerimpedances for obtaining optimal signal levels. This is not trivial asthe impedances of ultrasonic transducers most often dependent on thetemperature and differ among samples. Last but not least, such anapproach is more sensitive to electrical noise.

In recent years, new types of operational amplifiers have been designedthat make the very low impedances feasible, even in battery operatedflow meters. Especially, the so-called current feedback operationalamplifiers, which have a lower input impedance on the inverting inputterminal but also have a higher bandwidth and a lower power consumptionand a higher gain at high frequencies than other types of operationalamplifiers, are beneficial for use in the present invention.

FIG. 4 illustrates schematically a coupling of ultrasonic transducers inan ultrasonic flow meter according to another embodiment of theinvention, in which the same circuit is used as signal generator SG andas receiver circuit RC. The obvious advantage of this embodiment is thatonly one common operational amplifier OP and, consequently, also onlyone set of feedback components for providing the desired resultingfeedback impedance Zfb are needed.

The price to be paid for this cost saving is that a switching of thesignal way is needed during the transmission of the signal. As can beseen from FIG. 4, the two switches SW1, SW2, which connect the twotransducers TR1, TR2, respectively, to the combined signal generator andreceiver circuit SG/RC, each have three possible positions.

This means that each of the two transducers TR1, TR2 can be:

-   -   1. set up to be the transducer to transmit the ultrasonic signal        by being coupled to the output terminal of the operational        amplifier OP, from which it is to receive the electrical        transmission signal, the common circuit SG/RC being operated as        a signal generator,    -   2. set up to be the transducer to receive the ultrasonic signal        by being coupled to the inverting input terminal of the        operational amplifier OP, to which it is to transmit the        electrical reception signal, the common circuit SG/RC being        operated as a receiver circuit, or    -   3. disconnected from the common circuit SG/RC whenever the other        transducer is connected to the common circuit.

Thus, by setting and changing the positions of the switches SW1, SW2appropriately at the right times, the desired signal paths of theelectrical transmission signal, the ultrasonic signal and the electricalreception signal can be obtained. For transmitting an ultrasonic signalfrom the first transducer TR1 to the second transducer TR2, the firsttransducer TR1 is first set up to transmit the ultrasonic signal byconnecting it to the common circuit SG/RC being operated as a signalgenerator while the second transducer TR2 is disconnected from thecommon circuit SG/RC. Subsequently, when the ultrasonic signal has beentransmitted by the first transducer TR1 but before it reaches the secondtransducer TR2, the first transducer TR1 is disconnected from the commoncircuit SG/RC and the second transducer TR2 is set up to receive theultrasonic signal by connecting it to the common circuit SG/RC beingoperated as a receiver circuit. For transmitting the ultrasonic signalin the opposite direction, the connections of the two transducers TR1,TR2 is simply swapped as compared to the above description.

FIG. 5 illustrates schematically a coupling of ultrasonic transducers inan ultrasonic flow meter according to yet another embodiment of theinvention. In this case, there are two receiver circuits RC1, RC2, eachcomprising an operational amplifier OPrc1, OPrc2 and a negative feedbackcircuit with impedances Zfb,rc1 and Ffb,rc2, respectively. This allowsthe two transducers TR1, TR2 to transmit an ultrasound signalsimultaneously.

In order to do this, both of the transducers TR1, TR2 are firstconnected to the signal generator SG by setting the switches SW1, SW2 inthe appropriate positions. An electrical transmission signal istransmitted simultaneously to the two transducers TR1, TR2, from whichit is transmitted into the flow path (FP) as an ultrasonic signal fromeach of the transducers TR1, TR2. Before the ultrasonic signal from thefirst transducer TR1 reaches the second transducer TR2 and vice versa,the positions of the switches SW1, SW2 are changed so that each of thetransducers TR1, TR2 is connected to one of the receiver circuits RC1,RC2.

In this way, the ultrasonic signal can be sent both upstream anddownstream along the flow path FP in a single operation. However, inorder to level out any possible minor metering errors due to the factthat the two receiver circuits RC1, RC2 cannot be constructed to becompletely identical, it should be assured that for every transmissionof the ultrasonic signals, the connections between the transducers TR1,TR2 and the receiver circuits RC1, RC2 are interchanged, so that a giventransducer TR1, TR2 is only connected to the same receiver circuit RC1,RC2 every second time.

Another difference in the coupling shown in FIG. 5 as compared to thecouplings shown in the previous figures is that the digital pulsatingsignal is connected to the inverting input terminal of the operationalamplifier OPsg of the signal generator SG through a filter impedanceZfilt, which is of influence on the amplification and the filtration ofthe electrical transmission signal, while the non-inverting inputterminal of the operational amplifier OPsg is grounded.

Compared to the configuration of the signal generators SG shown in theprevious figures, in which the input common mode voltage of theoperational amplifier will exhibit some variation as the digitalpulsating signal varies, this configuration has the advantage that theinput common mode voltage is kept at a constant DC level. This is ofimportance for some types of operational amplifiers, especially thefastest ones with the highest bandwidth.

FIG. 6 shows a diagram of most of the essential electronic components inan ultrasonic flow meter according to an embodiment of the invention.

In this diagram, the two inputs marked IN1 and IN2 indicate the input oftwo digital pulsating signals, the two resistors R4 and R7 are there forgenerating a symmetric transmission signal to the signal generator SGfrom the two digital signals, and the capacitor C7 forms an AC couplingbetween the incoming transmission signal and the signal generator SG.

The two resistors R3 and R9 and the two capacitors C1 and C9 form alow-pass filter for the incoming transmission signal (corresponding toZfilt in FIG. 5).

OPsg is the operational amplifier of the signal generator SG, which notonly amplifies the incoming transmission signal, but also is importantfor adjusting the output impedance of the signal generator SG to be verylow, i.e. substantially zero.

The three resistors R1, R2 and R40 and the two capacitors C29 and C30together constitute the negative feedback impedance of the operationalamplifier OPsg (corresponding to Zfb,sg in FIGS. 3 and 5).

The three resistors R11, R41 and R45 together form a voltage dividerdefining the reference voltages on the non-inverting inputs of OPsg andOPrc. Because OPsg and OPrc are both configured as inverting amplifiers,the reference voltages are the same as the input common mode voltages onthe two operational amplifiers OPsg and OPrc, respectively. Thereference voltages to the two operational amplifiers OPsg and OPrc aredecoupled by the two capacitors C2 and C25.

V3 (corresponding to VCC in FIGS. 7a and 7b ) is the positive supplyvoltage for the circuit. In low-power consumption flow meters, such asbattery operated flow meters, V3 can advantageously be cut off with aswitch (not shown) most of the time, so that the circuit is operated ata very low duty cycle.

In the embodiment shown in FIG. 6, the switching unit SU comprises fourswitches implemented on a single CMOS die in an integrated circuit forcoupling of the transducers TR1, TR2 to the signal generator SG and thereceiver circuit RC. The pins IN1-IN4 of the switching unit SU eachcontrols one of the four switches connected between the lines D1-S1,D2-S2, D3-S3 and D4-S4, respectively, by asserting a high voltage. Inthe diagram shown in FIG. 6, the switches are configured to transmit asignal from transducer TR1 to transducer TR2. For a signal to betransmitted from transducer TR2 to transducer TR1, opposite voltagesmust be applied to IN1-IN4. Preferable, IN1-IN4 are controlled by amicrocontroller.

The two resistors R10 and R36 are small current limiting resistors. Thestability of the operational amplifiers OPsg and OPrc is increased bylimiting the capacitive load on the amplifiers OPsg, OPrc.

The two capacitors C8 and C15 provide an AC coupling of the signals toand from the transducers TR1, TR2. This allows the use of single supplyvoltage operational amplifiers OPsg, OPrc as the ones shown in FIG. 6without any DC voltage on the transducers TR1, TR2.

The two resistors R13 and R14 are bleeders for discharge of thetransducers TR1, TR2 in case a charge is produced thereon due topyroelectric effects or due to other circumstances.

The two ultrasonic transducers TR1 and TR2 are preferably constituted bypiezoelectric transducers.

OPrc is the operational amplifier of the receiver signal RC, whichproduces an amplified output signal OUT from the electrical receptionsignal from the transducers TR1, TR2, but also is important foradjusting the input impedance of the receiver circuit RC to be very low,i.e. substantially zero.

The resistor R44 and the capacitor C5 constitute a filtering of thesupply voltage for the operational amplifier OPrc of the receivercircuit RC.

The two resistors R5 and R6 and the capacitor C4 together constitute thenegative feedback impedance of the operational amplifier OPrc(corresponding to Zfb,rc in FIG. 3).

The two resistors R8 (corresponding to RCC in FIGS. 7a and 7b ) and R43are used for current sensing of the power supply currents to theoperational amplifier OPsg. In some embodiments of the invention, R43may be omitted for increased stability. If OPsg is chosen to be anoperational amplifier with low supply voltage rejection ratio, switches(not shown) are needed to short-circuit R8 and R43 during transit timemeasurements.

The two capacitors C6 and C33 have the purpose of decoupling the supplyvoltages to OPsg. If the values of these two capacitors are too high,the voltages across R8 and R43 do not properly reflect the power supplycurrents to the operational amplifier OPsg. If, on the other hand, thevalues of C6 and C33 are too low, the operational amplifier OPsg ispotentially unstable.

The two capacitors C13 and C14 and the two resistors R12 and R15 arecomponents needed to combine the two supply current signals SCSa, SCSbinto a single signal supply current signal SCS. R12 and R15 also definethe DC voltage level for following circuits of the flow meter, such asan Analogue-Digital converter.

V1 is a supply voltage needed to generate the DC voltage level for thecombining circuitry C13, C14, R12, R15. By careful selection ofcomponents, V3 may be reused in place of the separate supply voltage V1.

FIGS. 7a and 7b illustrate schematically the set-up for performing afirst and a second step, respectively, in obtaining supply currentsignal SCS−, SCS+ of an active component driving an ultrasonictransducer.

In the first step, which is illustrated in FIG. 7a , a first, shortdigital pulsating input signal DPSa is used as an input for a signalgenerator SG driving an ultrasonic transducer TR1, TR2 to which it maybe connected through a switching unit SU. The signal generator SG maycomprise a negative feedback coupled operational amplifier (OP; OPsg),as shown in the previous figure, or it may comprise another activecomponent, which is able to amplify the input signal DPSa and provide anoutput impedance of the signal generator SG, which is very low, i.e.substantially zero. This active component might, for instance, beconstituted by a digital circuit driving the transducer.

A current sensing resistor RCC (corresponding to R8 in FIG. 6) isarranged in series between the active component of the signal generatorSG and the positive voltage supply VCC (corresponding to V3 in FIG. 6)for this active component. Now, by monitoring the voltage across thiscurrent sensing resistor RCC, a first supply current signal SCSarepresentative of the current supplied to the active component from thepositive voltage supply can be obtained as illustrated on the right sideof FIG. 7 a.

It should be noted that when the input signal DPSa stops oscillating,the transducer will continue to be oscillating for some time, stilldragging some current from the positive voltage supply through theactive component. This is reflected in the first supply current signalSCSa, which comprises a higher number of oscillations than the firstinput signal DPSa, as is indicated in FIG. 7a , the latest part of thesignal DPSa showing that the self-oscillating transducer TR1, TR2performs a dampened oscillation.

As can also be seen from the first supply current signal SCSaillustrated in FIG. 7a , this signal is truncated in the sense that onlyone half part of each oscillation is comprised in the signal, the signalvalue being zero in the other half part of each oscillation. This is dueto the fact that, if the active component of the signal generator iscoupled to be a non-inverting amplifier, the positive voltage supply VCConly delivers the current to the active component when the voltage ofthe input signal DPSa is higher than the input common mode voltage ofthe active component, and if the active component is coupled to be aninverting amplifier, the positive voltage supply VCC only delivers thecurrent to the active component when the voltage of the input signalDPSa is lower than the input common mode voltage of the activecomponent.

Therefore, in order to obtain a second supply current signal SCSbcomprising the other half part of each oscillation, the measurement isrepeated with another digital pulsating input signal DPSb, which isidentical to the first input signal DPSa with the one exception that thepolarity of the signal has been inversed.

It should be noted that the two supply current signals SCSa, SCSb can beobtained simultaneously from the positive and the negative voltagesupplies of the active component of the signal generator SG,respectively, if a similar current sensing resistor (not shown in FIGS.7a and 7b but corresponding to R43 in FIG. 6) is arranged in seriesbetween the active component and the negative voltage supply (not shown)for the active component.

FIG. 7c illustrates an advantageous way of connecting a signal generatorSG and a receiver circuit RC for obtaining such supply current signalsSCSa, SCSb.

In this case, the active component OPrc of the receiver circuit RC isused for amplifying the supply current signals SCSa, SCSb. Theconnection between the signal generator SG and the receiver circuit RCconsists of a switch SWconn in series with at high pass filterconsisting of a capacitor Cconn and a resistor Rconn.

As the connection SWconn, Cconn, Rconn is attached to a summation pointat the inverse input terminal of the active component OPrc of thereceiver circuit RC, the supply current signals SCSa, SCSb do not affectthe function of the ultrasonic transducers TR1, TR2 in any way.

Furthermore, due to the transit time of the ultrasonic signal betweenthe two transducers TR1, TR2 in the flow path FP, the supply currentsignals SCSa, SCSb and the signal transmitted between the transducersTR1, TR2 will reach the receiver circuit RC at different times.

At high frequencies, non-idealities of the components potentiallyinfluence the signals, and the supply current signals SCSa, SCSb mayinfluence the signal received through the flow path FP and vice versa. Aremedy for minimizing this effect is to measure the supply currentsignals SCSa, SCSb and the ultrasonic signal at different times anddisconnect unused circuit parts from the input terminal of the activecomponent OPrc of the receiving circuit RC by switches SW2, SWconn.

If the signal generator SG is configured as a Class A amplifier, thecurrent drawn into the positive voltage supply pin is substantiallyconstant, and a current sensing resistor (R43) has to be arranged inseries with the negative voltage supply for a useful signal to beobtained.

By subtracting the two supply current signals SCSa, SCSb from each otheras illustrated in FIG. 8a , a subtraction supply current signal SCS− isobtained, from which the oscillation period Tscs of the dampenedoscillation of the transducer TR1, TR2 can easily be determined bymeasuring the time difference between two appropriately chosen zerocrossings of the signal SCS− as is also indicated in FIG. 8 a.

The relation between the oscillation period Tscs and the frequency f_(D)and the angular frequency γ_(D) of the dampened transducer oscillationis well-known:

$\begin{matrix}{{Tscs} = {\frac{1}{f_{D}} = \frac{1}{2\;\pi\;\omega_{D}}}} & \left( {{Equation}\mspace{14mu} 3} \right)\end{matrix}$

Furthermore, by adding the two supply current signals SCSa, SCSb to eachother as illustrated in FIG. 8b , an addition supply current signal SCS+is obtained, the envelope Escs of the decreasing part of which can bedetermined. This envelope Escs has the shape of an exponential curvewith respect to the time t, the mathematical formula describing theenvelope beingEscs=−ke ^(−αt)  (Equation 4)where k is a constant and α is the damping coefficient of the dampenedoscillation of the transducer TR1, TR2.

In principle, both Ω_(D) and α could be found from each of the measuredsupply current signals SCSa, SCSb alone. However, the two quantities canbe determined with much higher accuracy using the subtraction supplycurrent signal SCS− and the addition supply current signal SCS+ asillustrated in FIGS. 8a and 8 b.

The two quantities ω_(D) and α are very useful for characterizing thetransducer, being indicative of the condition of the transducer, suchas, for instance, whether it might be broken or whether there might besome air around a transducer, which is supposed to be surrounded bywater, etc.

FIG. 9 illustrates a well-known equivalence diagram of an ultrasonictransducer TR, comprising a parallel capacitor Cpar coupled in parallelwith a series connection of a series inductor Lser, a series capacitorCser and a series resistor Rser.

FIG. 10 illustrates some of the steps in the derivation of a simpleequivalence diagram of an ultrasound flow meter, which can be used forsimulating the signal chain through a flow meter according to theinvention.

The equivalence diagram in the first part of FIG. 10 illustrates how theflow meter can be equated by a system comprising two ultrasonictransducers TR1, TR2, wherein the first transducer TR1, upon which atransmission signal in the form of a voltage signal Vtr1 is impressed,transmits an ultrasound signal to the second transducer TR2, which inturn produces a reception signal in the form of a current signal Itr2.

For a given input signal to the signal generator of the flow meter, thevoltage signal Vtr1 impressed on the first transducer TR1 can be takento be the same for each transit time measurement due to the fact thatall components of the signal generator are the same for eachmeasurement. This also means that the impressed signal Vtr1 can becalculated from the input signal using a filter model of the signalgenerator, once this filter model has been determined once and for all,or it can be recorded by an analogue-digital converter before thesimulation procedure, which is described below.

Introducing the equivalence diagram from FIG. 9 for each of the twotransducers TR1, TR2 in the equivalence diagram of the flow meter shownin the first part of FIG. 10 results in the diagram shown in the secondpart of FIG. 10. Here, it is noted that, due to the fact the ultrasoundsignal transmitted by the first transducer TR1 is proportional to thecurrent signal Itr1 passing through the transducer TR1, the ultrasoundsignal reaching the second transducer TR2 can be equated by a voltagesignal Vtr2 being impressed on the transducer TR2, said voltage signalVtr2 being proportional to the current signal Itr1 as indicated in FIG.10 by the proportionality factor K1.

The parallel capacitors Cpar1, Cpar2 have no influence on the impressedvoltages Vtr1, Vtr2 across the series connections Lser1, Cser1, Rser1and Lser2, Cser2, Rser2, respectively, in the equivalence diagram in thesecond part of FIG. 10 and can, thus, be ignored. Therefore, theresulting equivalence diagram of the flow meter to be used forsimulating the signal chain through the flow meter is the one shown inthe third and final part of FIG. 10.

The relations between the impressed voltage signals Vtr1, Vtr2 and theresulting current signals Itr1, Itr2 can be found by well-knowndifferential equations:

$\begin{matrix}\begin{matrix}{{{Itr}\; 1} = \frac{{Vtr}\; 1}{{{sLser}\; 1} + \frac{1}{{sCser}\; 1} + {{Rser}\; 1}}} \\{= {\frac{1}{{Lser}\; 1}\frac{s}{s^{2} + {s\frac{{Rser}\; 1}{{Lser}\; 1}} + \frac{1}{{Lser}\; 1\;{Cser}\; 1}}{Vtr}\; 1}}\end{matrix} & \left( {{Equation}\mspace{14mu} 5} \right) \\\begin{matrix}{{{Itr}\; 2} = \frac{{Vtr}\; 2}{{{sLser}\; 2} + \frac{1}{{sCser}\; 2} + {{Rser}\; 2}}} \\{= \frac{K\; 1\;{Itr}\; 1}{{{sLser}\; 2} + \frac{1}{{sCser}\; 2} + {{Rser}\; 2}}} \\{= {\frac{K\; 1}{{Lser}\; 1\;{Lser}\; 2}\frac{s}{s^{2} + {s\frac{{Rser}\; 1}{{Lser}\; 1}} + \frac{1}{{Lser}\; 1\;{Cser}\; 1}}}} \\{\frac{s}{s^{2} + {s\frac{{Rser}\; 2}{{Lser}\; 2}} + \frac{1}{{Lser}\; 2\;{Cser}\; 2}}{Vtr}\; 1} \\{= {K\; 2\frac{s}{s^{2} + {2\;\alpha_{1}s} + \omega_{1}^{2}}\frac{s}{s^{2} + {2\;\alpha_{2}s} + \omega_{2}^{2}}{Vtr}\; 1}}\end{matrix} & \left( {{Equation}\mspace{14mu} 6} \right)\end{matrix}$α₁ and α₂ are the damping coefficients relating to the first ultrasonictransducer TR1 and the second ultrasonic transducer TR2, respectively,corresponding to the damping coefficients that can be found from theenvelope of the addition supply current signals SCS+, as describedabove.

ω₁ and ω₂ are the undampened angular oscillation frequencies of thefirst ultrasonic transducer TR1 and the second TR2 ultrasonictransducer, respectively. The relation between these undampened angularoscillation frequencies ω₁, ω₂ used in the simulation equations and thecorresponding dampened angular oscillation frequencies ω_(D1) and ω_(D2)found by measuring the time difference between two appropriately chosenzero crossings of the dampened oscillation in the subtraction supplycurrent signals SCS−, as described above, is as follows:ω₁=√{square root over (ω_(D1) ²+α₁ ²)}

ω2=√{square root over (ω_(D2) ²+α₂ ²)}  (Equation 7)

K2 is a proportionality factor, which can be calculated. However, likethe specific component values of Cser1, Lser1, Rser1, Cser2, Lser2,Rser2 of the equivalence diagram in the last part of FIG. 10, the valueof K2 is not needed for using Equation 6 for simulating the signal chainthrough the flow meter.

The last expression of Equation 6, which is a differential equation fora circuit comprising two second order oscillating circuits, can besimulated by means of well-known mathematical tools, such as forinstance the Runge-Kutta method.

FIG. 11 illustrates some of the differences and similarities between aphysical and a simulated signal chain through a flow meter according tothe invention.

In the simulated signal chain, the physical transducers TR1, TR2 and theloads related to them are modelled, for instance as already describedabove.

In the fully simulated signal chain in FIG. 11, the output from thephysical signal generator, which filters a digital input signal from asignal controller and works as a driver for the first transducer TR1 byamplifying the filtered input signal, is simulated by a signal functionbeing filtered using a filter model of the signal generator before beingused as input function for the simulated transducer model. In anotherembodiment of the simulated signal chain, the input function for thesimulated transducer model may be produced by recording the actualoutput from the a physical signal generator by an analogue-digitalconverter, thus obtaining a signal chain that is partly physical, partlya simulated model.

The reception of the electrical reception signal by the receiver circuitand the subsequent signal processing in the physical signal chain isreplaced by signal processing alone in the simulated signal chain, thissignal processing optionally including a model (not shown) of thereceiver circuit.

Thus, if the signal function in the simulated signal chain does, infact, correspond to the input signal from the signal controller in thephysical signal chain, and if the filter models of the signal generatorand (optionally) the receiver circuit and the transducer models areadequate, the only difference between the output of the final signalprocessing of the simulated signal chain and the output of the finalsignal processing of the physical signal chain will be the time delayt_(d) of the ultrasonic signal in the flow path FP, which is not a partof the simulated signal chain.

The simulated model response being substantially identical to thephysical flow meter response except for the time delay t_(d) of theultrasonic signal in the flow path FP and a possible amplificationfactor makes it possible to determine this time delay t_(d) veryprecisely by following a method like the one illustrated schematicallyin FIG. 12.

The first step in this method is to characterize the two transducersTR1, TR2 by determining characteristic quantities, such as the angularfrequency ω_(D) and the damping coefficient α of dampened oscillationsof the transducers TR1, TR2 as described above.

Secondly, by using the known angular oscillation frequency ω of thetransmission signal used in the flow meter, an equivalence model of thetransducers TR1, TR2 can be found using Equations 5-7, and a numericalsimulation model of the transducers TR1, TR2 and the electronic circuitsof the signal generator SG and the receiver circuit RC can beestablished.

Thirdly, the system can be simulated by entering the input signalfunction (or alternatively a sampled version of the physicaltransmission signal reaching the first transducer) into the numericalsimulation model, whereby the simulation model response, i.e. the outputsignal from the receiver circuit RC as it would be according to themodel, if there was no time delay in the transmission of the ultrasonicsignal between the two transducers TR1, TR2, can be found.

In the fourth step of the method, the physical flow meter response, i.e.the physical reception signal actually received by the receiver circuitRC, is recorded.

Finally, the absolute transit time can be calculated by determining thetime delay of the physical flow meter response as compared to thesimulation model response.

FIG. 13 illustrates one example of how such a calculation of theabsolute transit time may be performed according to the invention.

Following the above-described method, an input signal entered to thesystem results in a measured physical flow meter response with a certaindelay and in a simulation model response with substantially no delay. Asmentioned above, if the equivalence model of the transducers TR1, TR2 isadequate, the two response signals will be substantially identicalexcept for the time delay, which is illustrated in FIG. 13.

Now, the absolute transit time, i.e. the time delay between the twosignals, can be determined very precisely, for instance by finding afiltered envelope of each of the two signals and determining the timedifference between the two points, in which the filtered envelopes havereached 50% of their maximum value, respectively. This approach forfinding the absolute transit time is illustrated schematically in FIG.13.

FIG. 14 illustrates an equivalence diagram of a signal generator SG andan ultrasonic transducer TR connected thereto. Basically, the signalgenerator SG in FIG. 14 is of the type shown in FIG. 5 with a feedbackresistance Rfb,sg and a filter resistance Rfilt, and the ultrasonictransducer TR is represented by the equivalence diagram illustrated inFIG. 9.

The active component OPsg of the signal generator SG is equipped with acurrent sensing resistor RCC arranged in series with the positivevoltage supply VCC as described above. The ultrasound transducer TR isconnected to the signal generator SG through a switch SW and bypassed bya bleeder resistor Rbleed corresponding to R13 and R14 in FIG. 6.

With the diagram in FIG. 14 as the point of departure, another methodfor determining the absolute transit time can be described, which is asprecise as the method described above but even more efficient in theterms of calculations needed, and which characterizes the impulseresponse of the transducer system rather than characterizing thetransducers TR1, TR2 themselves as described above.

In this alternative method, a single pulse supply current signal SPSCS1,SPSCS2 is recorded for each of the ultrasonic transducers TR1, TR2 byconnecting the respective transducer TR1, TR2 to the output of thesignal generator SG and obtaining a signal corresponding to thesubtraction supply current signal SCS− as described above for the othermethod.

The difference from the previous method is that in this case, thedigital pulsating input signals DPSa, DPSb have been replaced by asingle pulse like the one illustrated in FIG. 15a , which result insingle pulse supply current signals SPSCS1, SPSCS2 for the twotransducers TR1, TR2, respectively, like illustrated in FIGS. 15b and 15c.

It is obvious that the first oscillation of each of the two single pulsesupply current signals SPSCS1, SPSCS2 are somewhat distorted. This isdue to the fact that not all of the current supplied by the activecomponent OPsg of the signal generator SG passes through the oscillationcircuits Lser, Cser, Rser of the equivalent of the ultrasonictransducers TR1, TR2, respectively. In order to find the single pulseresponses SPRTR1, SPRTR2 of these oscillating circuits, the currentsthrough the two resistances Rfb,sg and Rfilt and the capacitance Cparmust be removed from the single pulse supply current signals SPSCS1,SPSCS2.

The current through Rfb,sg, which is connected to virtual ground, caneasily be found by opening the switch SW shown in FIG. 14 and repeatingthe measurements without any ultrasound transducers TR1, TR2 connectedto the signal generator. This results in a single pulse supply currentsignal SPSCS0 as the one illustrated in FIG. 15d , which is proportionalto the single pulse SP in FIG. 15a due to the fact that Rfb,sg is anohmic resistance.

The current through Rfilt cannot be measured as easy as the currentthrough Rfb,sg. However, since Rfilt is an ohmic resistance connected toground in parallel to Rfb,sg, the current through Rfilt can easily becalculated from SPSCS0, when the ratio between Rfb,sg and Rfilt isknown, and the two current signals can be subtracted from each of thetwo single pulse supply current signals SPSCS1, SPSCS2 for the twoultrasonic transducers TR1, TR2.

As for the current through Cpar, it consists of a transient spikecoinciding with the leading edge of the single pulse SP and anothertransient spike of opposite polarity coinciding with the trailing edgeof the single pulse SP. These spikes are of so short duration comparedto the oscillation periods of the single pulse supply current signalsSPSCS1, SPSCS2 for the two ultrasound transducers TR1, TR2, that theycan easily be subtracted from each of the single pulse supply currentsignals SPSCS1, SPSCS2 by simple interpolation.

After subtracting the currents through Rfb,sg, Rfilt and Cpar from eachof the two single pulse supply currents signals SPSCS1, SPSCS2 for thetwo ultrasound transducers TR1, TR2 as described above, the calculatedsingle pulse responses SPRTR1, SPRTR2 of the two ultrasound transducersTR1, TR2, respectively, which may look like illustrated in FIGS. 16a and16b , respectively, have been obtained.

The calculated single pulse response SPRSYS for the complete ultrasoundtransducer system, which is illustrated in FIG. 16c , is found as aconvolution of the single pulse responses SPRTR1, SPRTR2 of the twoultrasound transducers TR1, TR2 and may look like illustrated in FIG. 16c.

By repeating the calculated single pulse response SPRSYS a number oftimes with a suitable delay, an emulated flow meter responsecorresponding to an input signal comprising a number of pulses can becalculated, which is very similar to the actually measured flow meterresponse except for the time delay of the latter due to the transit timebetween the two ultrasound transducers TR1, TR2. FIGS. 17a and 17billustrate an emulated flow meter response RESPem and a correspondingmeasured flow meter response RESPms, respectively.

The absolute transit times may be determined by comparing filteredenvelopes of the emulated flow meter response and the measures flowmeter response, respectively, as illustrated schematically in FIG. 13and described above, or they may be determined by using Fast FourierTransformation (FFT) as described here below.

In the time domain, the estimated flow meter response z′(t) can becalculated from the equation:z′(t)=y ₁(t)*y ₂(t)*x(t)  (Equation 8)where y₁(t) and y₂(t) are the calculated single pulse responses of thetwo ultrasonic transducers TR1, TR2, respectively, represented by thetwo signals SPSCS1 and SPSCS2 in FIGS. 16a and 16b , respectively, andx(t) is the input signal on the input terminal of the active componentOPsg of the signal generator SG.

It should be noted that in Equation 8 above, the symbol ‘*’ is used asan operator indicating a convolution of the signals surrounding it,which is not to be confused with the multiplication operation oftenindicated by the same symbol.

In the time domain, the relation between the estimated flow meterresponse z′(t) and the measured flow meter response z(t) is given byz(t)≈z′(t−t _(d))  (Equation 9)where t_(d) is the time delay of the ultrasonic signal in the flow pathFP.

This reflects that the measured flow meter response and the estimatedflow meter response are close to being identical with exception of thetime delay t_(d) of the former.

If the estimated response had been perfect this would mean that, in thefrequency domain, the magnitudes of the measured flow meter responseZ(s) and of the estimated flow meter response Z′(s) would be the samefor all frequencies as shown in FIG. 18 a.

Furthermore, the phase angle between Z(s) and Z′(s) would changelinearly with the frequency as indicated in FIG. 18b , the slope of theline in FIG. 18b being the group time delay, corresponding to the timedelay t_(d) of the ultrasonic signal in the flow path FP.

FIG. 19 shows an actual flow meter response RESPem estimated asdescribed above and the corresponding actually measured flow meterresponse RESPms depicted in the same graph for displaying the similarityof the two signals and the time delay there between.

The magnitudes |Z′(s)| and |Z(s)| of the two flow meter responses in thefrequency domain, corresponding to RESPem and RESPms, respectively, areshown in FIGS. 20a and 20b , respectively. Although similar, it is clearthat the two graphs are not completely identical, indicating that theestimated flow meter response RESPem deviates slightly from the measuredflow meter response RESPms even if the time delay is ignored.

The phase angles between Z(s) and Z′(s) for different frequencies areshown in the graph in FIG. 21a . In theory, this graph should correspondto the one shown in FIG. 18b , but it is obvious that this is not thecase.

Looking closer at FIG. 21a , however, it emerges that the two graphs arenot as different as it seems at the first glance. Whereas the graph inFIG. 18b is linear, the graph in FIG. 21a is characterised by agenerally saw-toothed pattern due to the fact that the phase angles inthis graph have been “wrapped” to fall within the range from −π rad to+π rad.

It is obvious that the saw-toothed pattern of the graph in FIG. 21a isoverlaid by a certain amount of noise, which is mainly due to smallmagnitudes of Z(s) and/or Z′(s) affecting the calculations at somefrequencies. However, the segment of the graph between the two dottedlines, corresponding to a frequency range with large magnitudes of Z(s)and Z′(s) as can be seen from FIGS. 20a and 20b , is very close to beinglinear. FIG. 21b shows an enlargement of this segment of the graph inFIG. 21a . It should be noted that the slope of the linear segment canbe calculated by finding the phase angle between Z(s) and Z′(s) for onlytwo different frequencies.

The slope at different frequencies of the graph segment in FIG. 21b ,corresponding to the time delay t_(d) of the ultrasonic signal in theflow path FP as described above, are shown in FIG. 22. The time unit onthe vertical axis of this graph corresponds to the oscillation period ofthe transmission signal RESPms. Thus, according to FIG. 22, the measuredflow meter response RESPms is delayed by 2.3 oscillation periods ascompared to the emulated flow meter response RESPem, which is in perfectagreement with the two signal curves shown in FIG. 19.

It should be noted that the variation of the calculated time delayt_(d), i.e. of the slope of the graph segment in FIG. 22 is only a fewpercent of the oscillation period, which means that the time delay t_(d)calculated by this method is very precise as compared to other methodsknown in the art.

In practice, the actual flow meter response RESPms is best measuredusing a transmission signal comprising a number of pulses, whereas theemulated flow meter response RESPem is easiest calculated throughappropriate digital filtering of the single pulse response SPRSYScalculated as described above. This has no influence on the resultsobtained by the method.

Using methods like the ones described above, the absolute transit times,corresponding to t₁ and t₂ in Equation 1, can be determinedindependently of the transducer parameters with very high absoluteaccuracy (down to about 100 nanoseconds for 1 MHz transducers, which issignificantly more precise than what is possible in all previously knownsystems).

Normally, flow meters according to the invention will perform flowmetering at regular time intervals, typically in the range between 0.1second and 5 seconds. However, it should be noted that, for instance inorder to extend the life time of a battery supplying the electricity fora flow meter, the characterization of the transducers or the transmittedsignal and the simulation of the flow meter system do not need to berepeated for every flow metering performed by the flow meter.

The transducer characteristics change slowly over time due to aging ofthe transducers TR1, TR2 and more spontaneously due to changes in thetemperature of the fluid in the flow path FP in which they are arranged.

Thus, new transducer characterizations or signal characterizations anddetermination of an updated simulation model of the flow meter systemfor use in the calculation of absolute transit times may advantageouslyby performed at regular predetermined time intervals and/or when achange of temperature above a certain predetermine limit is detected,the temperature change being indicated by a change in the calculatedtransit time due to the dependency of ultrasound speed on thetemperature of the medium in which the ultrasound propagates.

Due to the high costs (and the high power consumption) of very fastanalogue-digital converters, slower converters may advantageously beused in flow meters according to the invention. However, as iswell-known from the Nyquist theorem, if a signal is sampled at afrequency lower than twice the maximum frequency occurring in thesignal, the analogue signal cannot be reconstructed without a certaindistortion.

Thus, if a low sampling frequency is used for recording the electricalreception signal received by the receiver circuit of a flow meteraccording to the invention, the physical flow meter response signal willbe distorted. If, however, the simulation model response is subjected tothe same undersampling, a similar distortion of this signal will takeplace, and the two response signals can still be compared for finding avery precise measure of the absolute transit time as described above.

The well-known spectral consequences of undersampling a continuoussignal are illustrated schematically in FIGS. 23a -c.

FIG. 23a illustrates an example of the frequency spectrum of acontinuous signal, while FIG. 23b illustrates how undersampling (in thiscase with a sampling frequency fs of ⅚ of the signal frequency) resultsin the creation of an infinite number of aliases of the originalspectrum.

FIG. 23c illustrates schematically how the undersampled signal can bereconstructed by changing the sampling frequency to the Nyquist samplingfrequency fs₂ and filtering the signal with an FIR reconstructionfilter, the frequency band of which is indicated in the figure.

The distortion of the reconstructions of undersampled continuous signalsis illustrated schematically in FIGS. 24a-b and 25a -b.

FIG. 24a illustrates an example of a continuous signal, while FIG. 24billustrates the digital samples obtained by sampling the continuoussignal of FIG. 24a at a sampling frequency of ⅚ of the signal frequency.

FIG. 25a illustrates the reconstruction of the signal of FIG. 24a usinga wide band FIR reconstruction filter, whereas FIG. 25b illustrates thereconstruction of the same signal using a narrow band FIR reconstructionfilter.

From comparing the reconstructed signals shown in FIGS. 25a and 25b tothe original signal shown in FIG. 24a , it is clear that the use of anarrow band FIR reconstruction filter results in a more severedistortion of the signal than does the use of a wide band FIRreconstruction filter. The optimum band width of the FIR reconstructionfilter to be used depends partly on the width of the individual aliasesin the undersampled spectrum, partly on the amount of noise in thesignal.

FIG. 26 illustrates schematically a method for finding the amplitude andphase of an undersampled continuous signal without using any FIRreconstruction filter.

The upper half of FIG. 26 comprises two frames illustrating a continuoussignal and the digital samples obtained by sampling this continuoussignal at a sampling frequency of ⅚ of the signal frequency,respectively.

The relation between the signal frequency and the sampling frequencymeans that for each sixth oscillations of the continuous signal, fivesamples will be collected. If the continuous signal is stationary, thefive samples from a period of six oscillations will correspond exactlyto the five samples from the previous six oscillations and to the fivesamples from the following six oscillations.

If a relatively long input signal is used, the midmost part of thesignal can be considered to be substantially stationary as indicated inthe upper half of FIG. 26, and the samples obtained from this part ofthe signal can be sorted out and summed up in five groups, eachcontaining a number of “similar” samples from different periods of sixoscillations of the continuous signal as illustrated in the lower halfof FIG. 26.

If this sorting and summing up of samples is done in an appropriate way,these five groups of samples together form an “average sampling” of asingle oscillation, which corresponds to a single oscillation of thesubstantially stationary part of the continuous signal and from whichthe amplitude and phase of the continuous signal can be determined bymeans of Digital Fourier Transformation DFT.

As mentioned above, the difference between the transit times of twodifferent ultrasonic signals, corresponding to the quantity (t₁−t₂) inEquation 1, can easily be found by comparing the phases of thecorresponding two electrical reception signals received by the receivercircuit RC of the flow meter. Thus, in order to find this difference ina system using undersampling, the transmission signals shouldadvantageously be relatively long, assuring that there is enoughinformation in the samples from the substantially stationary part of thesignal to determine the phase of the signal with a sufficient accuracy.

FIG. 27 illustrates schematically a method for reconstructing anundersampled continuous signal without distortion.

Again, the continuous signal, which is shown in the top of FIG. 27, issampled with an analogue-digital converter working at a samplingfrequency of ⅚ of the signal frequency. However, in this method, thesignal is transmitted and sampled six times, the timing of the samplingbeing displaced corresponding to ⅕ of the oscillation period of thecontinuous signal, the resulting six sets of interleaved samples fromwhich samplings are illustrated schematically in the central part ofFIG. 27.

By combining the interleaved samples appropriately, samplescorresponding to a sampling frequency of 5 times the signal frequencyare obtained. Only two times the signal frequency needed (according tothe Nyquist theorem), this is more than sufficient to reconstruct thesignal without any distortion.

In order to determine precise values of the absolute transit times t₁and t₂ to be added together to find the quantity (t₁+t₂) in Equation 1,the transmission signals should advantageously be relatively sharp,short and well-defined.

Taking the above consideration into account, the digital signalprocessing on the physical and simulated model responses of a flow meteraccording to the invention in order to obtain a very precise value ofthe absolute transit time may be performed as illustrated schematicallyin FIG. 28.

First, if the signal is undersampled, upsampling and anti-aliasfiltering is performed in order to reconstruct the signal.

After that, an optional filtering including bandwidth limitation may beperformed in order to improve the signal-noise ratio of the signal.

The relatively sharp, short and well-defined transmission signal, whichis advantageous for obtaining a very precise absolute transit timedetermination, cf. the above, is emulated from the actual received andfiltered signal in two steps:

The first emulation step consists in obtaining an emulated substantiallystationary signal by adding to the received signal delayed versions ofthe signal itself. If, for instance, the transmitted signal containsfive oscillations, versions of the received and filtered signal, whichare delayed by five, ten, fifteen, etc. periods of the signal are addedto the actually received and filtered signal. Due to the completelinearity of the system, the principle of superposition assures that theresulting signal is exactly similar to the filtered version of thesignal that would have been received if the transmission signal hadcontained ten, fifteen, twenty, etc. oscillation periods.

The second emulation step consists in subtracting a delayed version ofthe emulated substantially stationary signal from the emulatedsubstantially stationary signal itself. If, for instance, the subtractedsignal is delayed by two periods of the signal, the principle ofsuperposition assures that the resulting signal is exactly similar tothe filtered version of the signal that would have been received if thetransmission signal had contained two oscillation periods. If thissubtraction had been done with two versions of the original received andfiltered signal corresponding to a transmission signal containing onlyfive oscillation periods, the resulting signal would be a signalcorresponding to a transmitted signal having two pulses followed by apause of three oscillation periods and then by another two pulsesopposite in phase from the first two pulses. Obviously, such an oddsignal would not be very suitable for the purpose.

Now, the envelope of the emulated short signal is calculated and thepoint of time, on which 50% of the maximum value of the envelope hasbeen reached, is found as illustrated in FIG. 13.

Finally, the absolute transit time is determined by subtracting thecorresponding point of time relating to the envelope of the simulationmodel response signal calculated from the transducer characteristics asdescribed above.

It should be noted that the scope of the invention is in no way to beunderstood as being limited to the above-described embodiments of theinvention, which are only to be seen as examples of a multitude ofembodiments falling within the scope of the invention as defined by thebelow patent claims.

LIST OF REFERENCE SYMBOLS

-   -   CC. Common conductor for signal generator and receiver circuit    -   Cconn. Capacitor in connection between signal generator power        supply and receiver circuit    -   Cpar. Parallel capacitor in equivalence diagram for ultrasonic        transducer    -   Cpar1. Parallel capacitor in equivalence diagram for first        ultrasonic transducer    -   Cpar2. Parallel capacitor in equivalence diagram for second        ultrasonic transducer    -   Cser. Series capacitor in equivalence diagram for ultrasonic        transducer    -   Cser1. Series capacitor in equivalence diagram for first        ultrasonic transducer    -   Cser2. Series capacitor in equivalence diagram for second        ultrasonic transducer    -   DFT. Digital Fourier Transformation    -   DPSa. First digital pulsating input signal    -   DPSb. Second digital pulsating input signal    -   Escs. Envelope of supply current signal    -   FP. Flow path for fluid flow    -   fs. Undersampling frequency    -   fs₂. Nyquist sampling frequency    -   Itr1. Current through first transducer in equivalence diagram    -   Itr2. Current through second transducer in equivalence diagram    -   Lser. Series inductor in equivalence diagram for ultrasonic        transducer    -   Lser1. Series inductor in equivalence diagram for first        ultrasonic transducer    -   Lser2. Series inductor in equivalence diagram for second        ultrasonic transducer    -   OP. Operational amplifier common for signal generator and        receiver circuit    -   OPrc. Operational amplifier in receiver circuit    -   OPrc1. Operational amplifier in first receiver circuit    -   OPrc2. Operational amplifier in second receiver circuit    -   OPsg. Operational amplifier in signal generator    -   Rbleed. Bleeder resistor for ultrasound transducer    -   RC. Receiver circuit    -   RCC. Current sensing resistor for power supply current    -   Rconn. Resistor in connection between signal generator power        supply and receiver circuit    -   RESPem. Emulated flow meter response    -   RESPms. Measured flow meter response    -   Rfb,sg. Feedback resistance in signal generator    -   Rfilt. Filter resistance    -   Rser. Series resistor in equivalence diagram for ultrasonic        transducer    -   Rser1. Series resistor in equivalence diagram for first        ultrasonic transducer    -   Rser2. Series resistor in equivalence diagram for second        ultrasonic transducer    -   SCSa. First half of supply current signal    -   SCSb. Second half of supply current signal    -   SCS−. Subtraction supply current signal    -   SCS+. Addition supply current signal    -   SG. Signal generator    -   SP. Single pulse    -   SPRSYS. Single pulse response of the system    -   SPRTR1. Single pulse response of first ultrasonic transducer    -   SPRTR2. Single pulse response of second ultrasonic transducer    -   SPSCS0. Single pulse supply current signal without any        ultrasonic transducers    -   SPSCS1. Single pulse supply current signal for first ultrasonic        transducer    -   SPSCS2. Single pulse supply current signal for second ultrasonic        transducer    -   SPU. Signal processing unit    -   SU. Switching unit    -   SW. Switch    -   SW1. First switch    -   SW2. Second switch    -   SWconn. Switch in connection between signal generator power        supply and receiver circuit    -   t_(d). Time delay of ultrasonic signal in flow path    -   Tscs. Oscillation period of supply current signal    -   TR1. First ultrasonic transducer    -   TR2. Second ultrasonic transducer    -   TR. Ultrasonic transducer    -   VCC. Positive power supply voltage    -   Vtr1. Voltage impressed on first transducer in equivalence        diagram    -   Vtr2. Voltage impressed on second transducer in equivalence        diagram    -   Zad. Adaptation impedance    -   Zfb. Feedback impedance in combined signal generator and        receiver circuit    -   Zfb,rc. Feedback impedance in receiver circuit    -   Zfb,rc1. Feedback impedance in first receiver circuit    -   Zfb,rc2. Feedback impedance in second receiver circuit    -   Zfb,sg. Feedback impedance in signal generator    -   Zfilt. Filter impedance    -   Zsig. Signal impedance    -   α₁. Damping coefficient relating to first ultrasonic transducer    -   α₂. Damping coefficient relating to second ultrasonic transducer    -   ω₁. Undampened angular oscillation frequency of first ultrasonic        transducer    -   ω₂. Undampened angular oscillation frequency of second        ultrasonic transducer    -   ω_(D1). Dampened angular oscillation frequency of first        ultrasonic transducer    -   ω_(D2). Dampened angular oscillation frequency of second        ultrasonic transducer

What is claimed is:
 1. An ultrasonic flow meter comprising at least oneultrasonic transducer and a signal generator for generating electricalsignals to the at least one transducer, the signal generator comprisingan active component, wherein the ultrasonic flow meter comprises acurrent measuring arrangement configured to measure at least one powersupply current to said active component of said signal generator toestablish at least one supply current signal, and wherein said currentmeasuring arrangement comprises an impedance inserted in series betweena source of a positive or negative supply voltage and the activecomponent.
 2. The ultrasonic flow meter of claim 1, wherein said currentmeasuring arrangement comprises a resistor inserted in series between asource of a positive supply voltage and the active component.
 3. Theultrasonic flow meter of claim 1, wherein said current measuringarrangement comprises a resistor inserted in series between a source ofa negative supply voltage and the active component.
 4. The ultrasonicflow meter of claim 1, wherein said ultrasonic flow meter is arranged tofeed a pulsating input signal to said signal generator to generate saidelectrical signal to the at least one transducer.
 5. The ultrasonic flowmeter of claim 1, wherein said ultrasonic flow meter is arranged to feeda single pulse input signal to said signal generator to generate saidelectrical signal to the at least one transducer.
 6. The ultrasonic flowmeter of claim 1, wherein said at least one transducer is at least twotransducers and wherein said ultrasonic flow meter is arranged toperform said measure at least one power supply current to said activecomponent of said signal generator at least one time for each of said atleast two transducers to establish at least one supply current signalfor each of said at least two transducers.
 7. The ultrasonic flow meterof claim 1, wherein said ultrasonic flow meter is arranged to calculatean emulated flow meter response on the basis of said at least one supplycurrent signal.
 8. The ultrasonic flow meter of claim 1, wherein saidultrasonic flow meter is arranged to calculate an emulated flow meterresponse on the basis of at least one transducer characterizing quantitydetermined from said at least one supply current signal.
 9. Theultrasonic flow meter of claim 8, wherein said at least one transducercharacterizing quantity determined from said at least one supply currentsignal comprises at least one of an oscillation period, and a dampingcoefficient.
 10. The ultrasonic flow meter of claim 9, wherein said atleast one transducer characterizing quantity is determined from adampened oscillation part of one or more of said at least one supplycurrent signals or a signal derived therefrom.
 11. The ultrasonic flowmeter of claim 1, wherein said at least one transducer is at least twotransducers and wherein said ultrasonic flow meter is a transit timebased ultrasonic flow meter arranged to use said established at leastone supply current signal together with a measured flow meter responseto determine an absolute transit time of an ultrasonic signal in a flowpath of said ultrasonic flow meter.
 12. The ultrasonic flow meter ofclaim 1, wherein said at least one transducer is at least twotransducers and wherein said ultrasonic flow meter is a transit timebased ultrasonic flow meter arranged to use a measured flow meterresponse together with an emulated flow meter response calculated fromsaid established at least one supply current signal to determine anabsolute transit time of an ultrasonic signal in a flow path of saidultrasonic flow meter.
 13. The ultrasonic flow meter of claim 1, whereinsaid at least one transducer is at least two transducers and whereinsaid ultrasonic flow meter is a transit time based ultrasonic flow meterarranged to use a measured flow meter response together with anequivalence model of said at least two transducers determined fromtransducer characterizing quantities determined from said established atleast one supply current signal to determine an absolute transit time ofan ultrasonic signal in a flow path of said ultrasonic flow meter. 14.The ultrasonic flow meter of claim 1, wherein said at least onetransducer is at least two transducers and wherein said ultrasonic flowmeter is a transit time based ultrasonic flow meter arranged tocalculate an indication of a flow of fluid in a flow path of saidultrasonic flow meter on the basis of at least one absolute transit timedetermined from said at least one supply current signal.
 15. Anultrasonic flow metering method for an ultrasonic flow meter comprisingat least one ultrasonic transducer and a signal generator with an activecomponent for generating electrical signals to the at least onetransducer, the ultrasonic flow metering method comprising the steps of:measuring at least one power supply current to said active component ofsaid signal generator; and establishing at least one supply currentsignal based on said measuring, wherein the ultrasonic flow meterincludes a current measuring arrangement that comprises an impedanceinserted in series between a source of a positive or negative supplyvoltage and the active component, and wherein the measuring measures viathe current measuring arrangement.
 16. The ultrasonic flow meteringmethod of claim 15, wherein said measuring at least one power supplycurrent comprises measuring a voltage across at least one currentsensing resistor arranged in series between said active component ofsaid signal generator and at least one voltage supply of said signalgenerator.
 17. The ultrasonic flow metering method of claim 15,comprising the step of feeding a pulsating input signal to said signalgenerator to generate said electrical signal to the at least onetransducer.
 18. The ultrasonic flow metering method of claim 15,comprising the step of feeding a single pulse input signal to saidsignal generator to generate said electrical signal to the at least onetransducer.
 19. The ultrasonic flow metering method of claim 15, whereinsaid at least one transducer is at least two transducers and whereinsaid ultrasonic flow metering method comprises the step of repeatingsaid step of measuring at least one power supply current to said activecomponent of said signal generator to establish at least one supplycurrent signal for each of said at least two transducers.
 20. Theultrasonic flow metering method of claim 15, comprising the step ofcalculating an emulated flow meter response on the basis of said atleast one supply current signal.
 21. The ultrasonic flow metering methodof claim 15, comprising the step of calculating an emulated flow meterresponse on the basis of at least one transducer characterizing quantitydetermined from said at least one supply current signal.
 22. Theultrasonic flow metering method of claim 21, wherein said at least onetransducer characterizing quantity determined from said at least onesupply current signal comprises at least one of an oscillation period,and a damping coefficient.
 23. The ultrasonic flow metering method ofclaim 22, wherein said at least one transducer characterizing quantityis determined from a dampened oscillation part of one or more of said atleast one supply current signals or a signal derived therefrom.
 24. Theultrasonic flow metering method of claim 15, wherein said at least onetransducer is at least two transducers and wherein said ultrasonic flowmeter is a transit time based ultrasonic flow meter, the ultrasonic flowmetering method comprising the step of measuring a flow meter response;and determining an absolute transit time of an ultrasonic signal in aflow path of said ultrasonic flow meter using said established at leastone supply current signal and said measured flow meter response.
 25. Theultrasonic flow metering method of claim 15, wherein said at least onetransducer is at least two transducers and wherein said ultrasonic flowmeter is a transit time based ultrasonic flow meter, the ultrasonic flowmetering method comprising the step of calculating and emulated flowmeter response from said established at least one supply current signal;measuring a flow meter response; and determining an absolute transittime of an ultrasonic signal in a flow path of said ultrasonic flowmeter using said emulated flow meter response and said measured flowmeter response.
 26. The ultrasonic flow metering method of claim 15,wherein said at least one transducer is at least two transducers andwherein said ultrasonic flow meter is a transit time based ultrasonicflow meter, the ultrasonic flow metering method comprising the step ofdetermining at least one transducer characterizing quantity from said atleast one supply current signal for each of said at least twotransducers; determining an equivalence model of said at least twotransducers from said at least one transducer characterizing quantityfor each of said at least two transducers; measuring a flow meterresponse; and determining an absolute transit time of an ultrasonicsignal in a flow path of said ultrasonic flow meter using saidequivalence model and said measured flow meter response.
 27. Theultrasonic flow metering method of claim 15, wherein said at least onetransducer is at least two transducers and wherein said ultrasonic flowmeter is a transit time based ultrasonic flow meter, the ultrasonic flowmetering method comprising the step of measuring at least one flow meterresponse; determining at least one absolute transit time using said atleast one supply current signal and said at least one measured flowmeter response; and calculating an indication of a flow of fluid in aflow path of said ultrasonic flow meter using said at least one absolutetransit time.